Partial response receiver and related method

ABSTRACT

A multi-phase partial response equalizer circuit includes sampler circuits that sample an input signal to generate sampled signals in response to sampling clock signals having different phases. A first multiplexer circuit selects one of the sampled signals as a first sampled bit to represent the input signal. A first storage circuit coupled to an output of the first multiplexer circuit stores the first sampled bit in response to a first clock signal. A second multiplexer circuit selects one of the sampled signals as a second sampled bit to represent the input signal based on the first sampled bit. A second storage circuit stores a sampled bit selected from the sampled signals in response to a second clock signal. A time period between the second storage circuit storing a sampled bit and the first storage circuit storing the first sampled bit is substantially greater than a unit interval in the input signal.

CROSS REFERENCES TO RELATED APPLICATIONS

This patent application is a continuation of U.S. patent application Ser. No. 12/835,500, filed Jul. 13, 2010, which claims the benefit of U.S. provisional patent application No. 61/228,348, filed Jul. 24, 2009, both of which are incorporated by reference herein in their entireties.

BACKGROUND

The present disclosure relates to electronic circuits, and more particularly, to a partial response receiver and related methods.

In digital systems, data is transmitted as a sequence of level-encoded electrical pulses (i.e., bits) through a signaling path. Electrical pulses transmitted on a band-limited signaling path disperse in time as they travel from a transmitter to a receiver. Time-domain dispersion results in a blending of neighboring pulses. This effect is known as dispersion-type inter-symbol interference (ISI). Dispersion-type ISI becomes more pronounced at faster signaling rates, ultimately degrading the signal quality to the point at which distinctions between originally transmitted signal levels may be lost.

A partial response equalizer (PRE) can be used in a receiver to mitigate dispersion-type ISI that results from transmitting data through a signaling path. A PRE compares each bit in an incoming data signal to multiple threshold levels. The threshold levels are selected based on possible digital sequences of one or more previously received bits in the incoming data signal.

FIG. 1 illustrates a prior art double data rate (DDR) partial response equalizer 100 that samples two bits in an incoming data signal during each period of a sampling clock signal. Sampler circuits 101A-101D compare the incoming data signal V_(IN) to thresholds +/−α along parallel sampling paths to generate sampled bits DPE, DNE, DPO, and DNO, respectively. Sampler circuits 101A-101B are driven by clock signal CLK, and sampler circuits 101C-101D are driven by clock signal CLKB. CLK and CLKB are driven out of phase. Each of multiplexers 104A-104B selects one of the sampled bits based on a previously sampled bit, DO and DE, respectively. The sampled bits selected by multiplexers 104A-104B are stored in latches 105A-105B in response to CLKB and CLK as previously sampled bits DE and DO, respectively.

The delay of the longest feedback path in PRE 100 is shown below in equation (1).

T _(C,Q) +T _(MUX,SO) +T _(SU)<1 (UI)   (1)

In equation (1), T_(C,Q) is the clock-to-Q output delay of each of latches 105A and 105B, T_(MUX,SO) is the select-to-output delay of each of multiplexers 104A and 104B, T_(SU) is the setup delay of each of latches 105A and 105B, and UI is one unit interval (i.e., bit period) in V_(IN). Sampling V_(IN) along parallel paths in circuit 100 does not reduce the longest feedback path delay. PRE 100 imposes a lower limit on each unit interval as shown in equation (1). The lower limit on each unit interval may make it difficult to employ conventional partial response designs and their associated benefits as clock rates continue to increase. Therefore, it would be desirable to provide a partial response equalizer that performs the feedback selection in less time.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a prior art partial response equalizer that samples two bits in an incoming data signal during each period of a sampling clock signal.

FIG. 2 illustrates a system that includes a transmitter on a first integrated circuit that transmits signals through a signal path to a receiver on a second integrated circuit.

FIG. 3A illustrates a block diagram of a partial response equalizer (PRE) circuit that uses a time borrowing technique and that generates each sampled bit using the value of one previously sampled bit.

FIG. 3B is a flow chart that illustrates a process performed by the PRE circuit of FIG. 3A.

FIG. 3C is a timing diagram that illustrates example waveforms for some of the signals in the PRE circuit of FIG. 3A.

FIG. 4 illustrates an example of a PRE circuit that uses a time borrowing technique and that generates each sampled bit based on the values of two previously sampled bits.

FIG. 5A illustrates an example of a PRE circuit that uses a time borrowing technique and that generates each sampled bit based on the values of three previously sampled bits.

FIG. 5B illustrates examples of sense amplifiers in the PRE circuit of FIG. 5A.

FIG. 5C illustrates examples of flip-flops and multiplexers in the PRE circuit of FIG. 5A.

FIG. 5D is a timing diagram that illustrates example waveforms for the clock signals of FIG. 5A.

FIG. 6A illustrates another example of a PRE circuit that uses a time borrowing technique and that generates each sampled bit based on the values of three previously sampled bits.

FIG. 6B is a timing diagram that illustrates example waveforms for the clock signals of FIG. 6A.

DETAILED DESCRIPTION

A multi-data rate PRE circuit allows extra time for signals to propagate through the PRE circuit between the time when a first sampled bit is stored and the time when a second sampled bit is stored without changing the periods of the clock signals that clock the storage circuits. The time between when the second sampled bit is stored and when the first sampled bit is stored is reduced. This technique reduces a timing constraint and allows the PRE circuit to function at a higher data rate.

A multi-data rate PRE circuit has multiplexers that select sampled bits from among sampled signals generated by sampler circuits based on previously selected values of the sampled bits. A first multiplexer selects a first sampled bit in response to a previously selected sampled bit that is transmitted from an output of a storage circuit. A second multiplexer selects a second sampled bit in response to the first sampled bit without storing the first sampled bit in a storage circuit in the path between the output of the first multiplexer and the select input of the second multiplexer.

FIG. 2 illustrates a system 200 that transmits data from one integrated circuit to another integrated circuit. System 200 includes a transmitter (TX) circuit 203 on integrated circuit 201 and a receiver (RX) circuit 204 on integrated circuit 202. Transmitter circuit 203 drives a data signal to receiver circuit 204. The data signal driven by transmitter circuit 203 is transmitted from the output terminal of transmitter circuit 203 through output pin 206, signal path 205, and input pin 207 to an input terminal of receiver circuit 204. Receiver circuit 204 samples the data signal to generate sampled bits RX Data that are representative of bits in the data signal in response to a sampling clock signal CLK. Receiver circuit 204 drives the sampled bits to other circuitry (not shown). Pin 206 is an output pin on integrated circuit 201, and pin 207 is an input pin on integrated circuit 202. Signal path 205 is outside integrated circuits 201 and 202.

The data signal may pick up inter-symbol interference (ISI) while being transmitted from transmitter 203 to receiver 204. A partial response equalizer (PRE) circuit is located in receiver circuit 204 on integrated circuit 202. Although not separately depicted in FIG. 2, the PRE circuit is described with reference to FIG. 3A.

FIG. 3A illustrates a block diagram of a partial response equalizer (PRE) circuit 300 that uses a time borrowing technique and that generates each sampled bit using the value of one previously sampled bit. PRE circuit 300 is referred to as a 1-tap PRE circuit. PRE circuit 300 samples an input signal that has a double data rate with respect to the sampling clock signal.

Input voltage signal V_(IN) often picks up inter-symbol interference (ISI) while being transmitted from a transmitter to a receiver. ISI is a degradation of a signal that is caused when one bit in a bit stream interferes with one or more adjacent bits in the bit stream. A partial response equalizer circuit generates multiple samples of a current bit in an input signal using different ISI possibilities from one or more previous bits, to produce multiple conditionally generated signal samples. The PRE uses the multiple signal samples to proceed with sampling of the current bit and then, as the earlier bit is resolved, the PRE circuit selects the appropriate sampled value (i.e., based on the proper threshold given the value of the earlier bit) and discards the other conditional samples. The operating speed of the PRE most often depends on the time it takes for the equalizer to determine the value of a bit in the input signal before the equalizer can use the value of that bit to determine the value of the next bit in the input signal.

PRE circuit 300 samples input voltage signal V_(IN) using two clock signals CLK and CLKB to generate sampled signals. Clock signals CLK and CLKB are 180° out of phase with each other. PRE circuit 300 then selects one of the sampled signals as a sampled bit that represents a bit in the input signal V_(IN) for each unit interval. Each unit interval is a bit period in the input signal V_(IN). Two unit intervals occur in each clock period of clock signals CLK and CLKB. V_(IN) has a double data rate with respect to CLK (or CLKB). Sampled bits DE and DO represent two parallel data streams (“even” and “odd”) that are each generated at a single data rate.

PRE circuit 300 includes sampler circuits 301A-301B, multiplexer circuits 302A-302B, and flip-flops 303A-303B. FIG. 3B is a flow chart that illustrates a process performed by PRE 300 in steps 311-316.

Sampler circuit 301A samples input signal V_(IN) in response to clock signal CLK to generate sampled signals DPE and DNE at step 311 shown in FIG. 3B. Sampler circuit 301B samples input signal V_(IN) in response to clock signal CLKB to generate sampled signals DPO and DNO at step 312. The DPE and DPO signals are conditionally generated assuming the immediately prior bit to have been a “logic 1,” and the DNE and DNO signals are conditionally generated assuming the immediately prior bit to have been a “logic 0.”

Multiplexer 302A then selects either DPE or DNE as a sampled bit DES in each clock period of CLK in response to a previously sampled bit DO at step 313. Sampled bit DES represents a bit in the input signal V_(IN). Multiplexer 302B selects either DPO or DNO as sampled bit DOS in each clock period of CLKB in response to sampled bit DES at step 314. Sampled bit DOS represents a bit in the input signal V_(IN) that is received by PRE 300 after DE. Because each of the sampled bits DES and DOS is selected based on the value of one previously sampled bit, PRE 300 is a 1-tap PRE circuit.

Flip-flop 303A stores the value of sampled bit DES as stored sampled bit DE in response to clock signal SCLK2 at step 315. Flip-flop 303B stores the value of sampled bit DOS as stored sampled bit DO in response to clock signal SCLK1 at step 316.

The output signal DO of flip-flop 303B is transmitted from the output of flip-flop 303B to the select input of multiplexer 302A. The output signal DES of multiplexer 302A is transmitted directly to the select input of multiplexer 302B without being stored in a storage circuit in the path between the output of multiplexer 302A and the select input of multiplexer 302B. This configuration of PRE circuit 300 allows the delay between flip-flop 303B storing an updated value of DO and flip-flop 303A storing an updated value of DE to be increased, and the delay between flip-flop 303A storing an updated value of DE and flip-flop 303B storing an updated value of DO to be decreased, without changing the duration of the periods of SCLK1 and SCLK2.

FIG. 3C illustrates example waveforms for signals CLK, CLKB, SCLK1, SCLK2, DE, and DO. As shown in FIG. 3C, clock signal SCLK1 is delayed by a delay of t_(D) relative to the phase of clock signal CLKB. The falling edge of clock signal SCLK2 is delayed relative to the rising edge of clock signal SCLK1 by a delay of γ. In the example of FIG. 3C, each of the clock signals CLK, CLKB, SCLK1, and SCLK2 has a 50% duty cycle. In alternative embodiments, clock signals CLK, CLKB, SCLK1, and SCLK2 have duty cycles that are greater than or less than 50%. There are many different techniques for generating clock signals SCLK1 and SCLK2. For example, SCLK1 and SCLK2 can be generated using one or more delay circuits (e.g., one or more inverters) that delay CLK or CLKB to generate SCLK1 and SCLK2. Clock signals SCLK1 and SCLK2 can also be generated, for example, by a delay-locked loop, a phase mixer, or a phase-locked loop.

Between each rising edge in SCLK1 and the next rising edge in SCLK2, a clock-to-Q output delay T_(C,Q) occurs in flip-flop 303B, then a select-to-output delay T_(MUX,SO) occurs in multiplexer 302A, and then a setup delay T_(SU) occurs in flip-flop 303A, as shown, for example, at the bottom of FIG. 3C. The delay between each rising edge in SCLK1 and the next rising edge in SCLK2 equals one unit interval (UI) plus γ. The delay between each rising edge in SCLK1 and the next rising edge in SCLK2 in PRE circuit 300 is constrained by equation (2).

T _(C,Q) +T _(MUX,SO) +T _(SU)<1 (UI)+γ  (2)

As discussed above, each rising edge in SCLK1 causes flip-flop 303B to store the value of DOS as sampled bit DO, and each rising edge in SCLK2 causes flip-flop 303A to store the value of DES as sampled bit DE. The delay between each rising edge in SCLK1 and the next rising edge of SCLK2 is increased by γ to provide additional time to complete the delay T_(C,Q)+T_(MUX,SO)+T_(SU) through flip-flop 303B, multiplexer 302A, and flip-flop 303A. The delay between each rising edge of SCLK2 and the next rising edge of SCLK1 is reduced by γ. The relative phases of SCLK1 and SCLK2 can be shifted by γ without changing the duration of the periods of SCLK1 and SCLK2. This technique is an example of time borrowing.

Time borrowing refers to a technique in which the time period to sample a first bit in an input data signal is increased by a particular amount so that the time to sample the first bit is greater than one unit interval. The time period to sample a second bit in the input data signal is decreased by the same amount so that the time period to sample the second bit is less than one unit interval. Time borrowing permits the circuitry of FIG. 3A and the other embodiments described herein to be used with a clock signal having a greater frequency than in prior art techniques, because the longest signal path has more time to complete as shown, for example, in equation (2) with respect to PRE circuit 300. Time borrowing can be used to extend the use of a variety of partial response equalizer techniques as clock signal speeds increase.

In one embodiment, the delay γ in PRE circuit 300 is equal to T_(F)/2, where T_(F)=T_(C,Q)+T_(SU). In this embodiment, the timing constraint of equation (2) becomes T_(MUX,SO)+T_(F)/2<1 (UI), which is less than equation (1) by T_(F)/2.

The longest signal path through PRE circuit 300 is the path from flip-flop 303B, through multiplexers 302A-302B, and back to flip-flop 303B. In PRE circuit 300, the sum of the clock-to-Q delay T_(C,Q) of flip-flop 303B, the select-to-out delay T_(MUX,SO) of multiplexer 302A, the select-to-out delay T_(MUX,SO) of multiplexer 302B, and the setup delay T_(SU) of flip-flop 303B is less than two unit intervals (UI), as shown in equation (3) below, where N=2 in PRE circuit 300. The longest signal path of PRE circuit 300 is less than or equal to one clock period of SCLK1. The constraint shown in equation (3) is not as limiting as the first constraint shown in equation (2) as long as γ<T_(F)/2.

T _(C,Q) +N·(T _(MUX,SO))+T _(SU) <N·(UI)   (3)

FIG. 4 illustrates an example of a partial response equalizer circuit 400 that uses a time borrowing technique and that generates each sampled bit based on the values of two previously sampled bits. PRE circuit 400 is a 2-tap PRE circuit. PRE circuit 400 samples an input signal that has a double data rate (DDR) with respect to the sampling clock signal. Partial response equalizer (PRE) circuit 400 can be operated at significantly higher data rates than previously known 2-tap DDR PRE circuits.

PRE circuit 400 receives an input voltage signal V_(IN) that represents an incoming data signal. PRE circuit 400 generates sampled bits from input signal V_(IN) using a time borrowing technique. PRE circuit 400 samples input voltage signal V_(IN) using clock signals CLK and CLKB to generate sampled signals, selects two of the sampled signals as sampled bits that represent input signal V_(IN) in two unit intervals, and stores the sampled bits in storage circuits. Each sampled bit is selected based on the values of two previously sampled bits.

The time intervals between successive rising edges of the clock signals SCLK1 and SCLK2 that drive the storage circuits are substantially different from a unit interval. The delays between a unit interval and the rising edges of SCLK1 and SCLK2 have been inserted intentionally and are not simply the result of clock skews that routing commonly causes in digital integrated circuits.

PRE circuit 400 generates one sampled bit DE (data even) that equals the logic state of the bit in the input signal V_(IN) for each even numbered unit interval. PRE circuit 400 generates one sampled bit DO (data odd) that equals the logic state of the bit in the input signal V_(IN) for each odd numbered unit interval. Two unit intervals in V_(IN) occur in each clock period of clock signals CLK and CLKB, and each unit interval corresponds to one bit period in V_(IN). PRE circuit 400 generates two sampled bits of data (DE and DO) that represent the input signal V_(IN) in two unit intervals. V_(IN) has a double data rate with respect to CLK (or CLKB).

PRE circuit 400 includes sense amplifier circuits 401A-401D, set-reset (SR) flip-flops 402A-402H, multiplexers 403A-403F, flip-flops 404A-404B, clock buffer 405, and inverting clock buffer 406. Clock signal CLK is transmitted to inputs of flip-flops 402A-402D, and clock signal CLKB is transmitted to inputs of flip-flops 402E-402H. CLK and CLKB are driven 180° out of phase with each other.

The input voltage signal V_(IN) is transmitted to a first input terminal of each of the sense amplifier circuits 401A-401D. Sense amplifier circuits 401A-401D compare V_(IN) to threshold voltage signals α₁, α₂, α₃, and α₄ to generate sampled signals that are provided to inputs of flip-flops 402A and 402H, 402B and 402G, 402C and 402F, 402D and 402E, respectively. The values of α₁, α₂, α₃, and α₄ represent magnitudes of the partial response to the preceding bits in input signal V_(IN) caused by ISI, i.e., assuming the preceding bits to have been 00, 01, 10, and 11, respectively.

Sense amplifier circuits 401A-401D and flip-flops 402A-402D are sampler circuits that sample V_(IN) at the rising edge of CLK. Sense amplifier circuits 401A-401D and flip-flops 402E-402H are sampler circuits that sample V_(IN) at the rising edge of CLKB. While there are many variations, FIG. 4 depicts one possible operation of the sampler circuits.

The sampled signals generated by sense amplifier circuits 401A-401D are stored in flip-flops 402A-402D, respectively, when CLK is in a logic high state. The sampled signals generated by sense amplifier circuits 401A-401D are stored in flip-flops 402H, 402G, 402F, and 402E, respectively, when CLKB is in a logic high state.

Multiplexer 403A selects one of the sampled signals stored in flip-flops 402A-402B based on the logic state of previously sampled bit DE, and multiplexer 403B selects one of the sampled signals stored in flip-flops 402C-402D based on the logic state of DE. Multiplexer 403C selects one of the output signals of multiplexers 403A-403B as sampled bit DES based on the logic state of previously sampled bit DO. Flip-flop 404A stores the output signal DES of multiplexer 403C as sampled bit DE in response to a rising edge in clock signal SCLK2.

Multiplexer 403D selects one of the sampled signals stored in flip-flops 402E-402F based on the logic state of sampled bit DO, and multiplexer 403E selects one of the sampled signals stored in flip-flops 402G-402H based on the logic state of DO. Multiplexer 403F selects one of the output signals of multiplexers 403D-403E as sampled bit DOS based on the logic state of DES. Flip-flop 404B stores the output signal DOS of multiplexer 403F as sampled bit DO in response to a rising edge in clock signal SCLK1. Because each sampled bit DES and DOS is selected based on the values of two previously sampled bits, PRE circuit 400 is a 2-tap PRE circuit.

The example waveforms shown in FIG. 3C can also be used to illustrate the operation of FIG. 4. Clock buffer 405 delays clock signal CLKB to generate clock signal SCLK1. SCLK1 is delayed with respect to CLKB by a delay t_(D) as shown, for example, in FIG. 3C. Inverting clock buffer 406 delays and inverts clock signal SCLK1 to generate clock signal SCLK2. Inverting clock buffer 406 delays inverted SCLK2 by a delay of γ relative to SCLK1 as shown, for example, in FIG. 3C.

PRE circuit 400 uses time borrowing. The delay between each rising edge in SCLK1 and the next rising edge in SCLK2 is constrained by equation (2) shown above. The delay between each rising edge in SCLK1 that causes flip-flop 404B to update DO and the next rising edge of SCLK2 that causes flip-flop 404A to update DE is increased by γ to equal (1 UI+γ). The delay γ provides additional time to complete the delay T_(C,Q)+T_(MUX,SO)+T_(SU) through flip-flop 404B, multiplexer 403C, and flip-flop 404A. The delay between each rising edge of SCLK2 and the next rising edge of SCLK1 is reduced by γ to equal (γ−1 UI) without adversely affecting the operation of PRE circuit 400, because the output signal DES of multiplexer 403C is transmitted directly to the select input of multiplexer 403F. The delay can be substantial relative to the duration of a unit interval, i.e., the delay γ can be 10%, 15%, 20%, 25%, 30%, or more of a unit interval. In this example, the delay between each rising edge of SCLK1 and the next rising edge of SCLK2 is 110%, 115%, 120%, 125%, 130%, or more of a unit interval, and the delay between each rising edge of SCLK2 and the next rising edge of SCLK1 is 90%, 85%, 80%, 75%, 70%, or less of a unit interval, respectively.

In one embodiment, the delay γ in PRE circuit 400 is equal to T_(F)/2, where T_(F)=T_(C,Q)+T_(SU). In this embodiment, the timing constraint of equation (2) becomes T_(MUX)+T_(F)/2<1 (UI) in PRE circuit 400, which provides a substantial reduction in this timing constraint relative to an implementation of PRE circuit 400 in which clock signals SCLK1 and SCLK2 are 180° out of phase.

After each rising edge in SCLK2, flip-flop 404A updates the value of DE based on the new value of DES. Then, multiplexers 403A-403B, 403C, and 403F update signals DES and DOS based on the updated value of DE. However, the next rising edge in SCLK1 causes flip-flop 404B to capture the value of DOS as sampled bit DO before multiplexer 403F updates DOS based on the value of DE that was updated in response to the preceding rising edge of SCLK2. The delay γ can be increased to reduce the possibility of a hold time violation in flip-flop 404B.

The longest signal path through PRE circuit 400 is the path from flip-flop 404B, through multiplexers 403C and 403F, and back to flip-flop 404B. In PRE circuit 400, the sum of the clock-to-Q delay T_(C,Q) of flip-flop 404B, the select-to-out delay T_(MUX,SO) of multiplexer 403C, the select-to-out delay T_(MUX,SO) of multiplexer 403F, and the setup delay T_(SU) of flip-flop 404B is less than two unit intervals (UI), as shown in equation (3) above, where N=2.

In an alternative embodiment of PRE circuit 400, non-sequential logic circuitry (e.g., inverters) is coupled between the output of multiplexer 403C and the select input of multiplexer 403F. The logic circuitry drives signal DES to the select input of multiplexer 403F.

FIG. 5A illustrates an example of a partial response equalizer (PRE) circuit 500 that uses a time borrowing technique and that generates each sampled bit based on the values of three previously sampled bits. PRE circuit 500 is a 3-tap PRE circuit. PRE circuit 500 samples an input signal that has a quad data rate (QDR) with respect to the sampling clock signal.

PRE circuit 500 receives an input voltage signal V_(IN) that represents an incoming data signal. PRE circuit 500 generates sampled bits from input signal V_(IN) using time borrowing. PRE circuit 500 samples input signal V_(IN) using clock signals C0, C1, C2, and C3 to generate sampled signals, selects 4 of the sampled signals as sampled bits D1-D4 that represent input signal V_(IN) in 4 unit intervals, and stores the sampled bits in storage circuits. Each sampled bit is selected based on the values of 3 previously sampled bits.

Four unit intervals occur in each clock period of clock signals C0-C3 in PRE circuit 500. Each unit interval corresponds to a bit period in V_(IN). V_(IN) has a quad data rate (QDR) with respect to sampling clock signals C0-C3. The techniques described herein can be extended to include PRE circuits that achieve virtually any data rate, including an octal data rate (i.e., 8 bits per sampling clock cycle), a decade data rate (i.e., 10 bits per sampling clock cycle), etc.

PRE circuit 500 includes 8 sense amplifier circuits within circuit block 501, 8 SR flip-flops within each of the circuit blocks 502-505, 4 multiplexers within each of the circuit blocks 511-514, multiplexers 521-532, flip-flops 541-544, and clock buffers 551-553. The sense amplifiers in circuit block 501 are shown in detail in FIG. 5B. As shown in FIG. 5B, circuit block 501 includes 8 sense amplifiers 501A-501H. FIG. 5C illustrates 8 SR flip-flops 561-568 and 4 multiplexers 571-574. Each of the circuit blocks 502-505 includes SR flip-flops 561-568 coupled as shown in FIG. 5C. Each of the circuit blocks 511-514 includes multiplexers 571-574 coupled as shown in FIG. 5C.

The input voltage signal V_(IN) is transmitted to input terminals of sense amplifier circuits 501A-501H as shown in FIG. 5B. Sense amplifier circuits 501A-501H compare V_(IN) to threshold voltage signals α₁, α₂, α₃, α₄, α₅, α₆, α₇,and α₈ to generate sampled signals S1-S8 that are provided to inputs of each of the circuit blocks 502-505. The values of α₁-α₈ represent magnitudes of the partial response to the preceding bits in input signal V_(IN) caused by ISI, i.e., assuming the preceding bits to have been 000, 001, 010, 011, 100, 101, 110, and 111, respectively.

Sense amplifier circuits 501A-501H and the SR flip-flops in circuit blocks 502-505 sample V_(IN) to generate sampled signals. Circuit blocks 502-505 each have 8 SR flip-flops 561-568 that store the sampled signals generated by sense amplifiers 501A-501H as 8 sampled signals Q1-Q8 in response to rising edges in clock signals C0, C1, C2, and C3, respectively. Clock signals C0, C1, C2, and C3 have relative phases of 0°, 90°, 180°, and 270°, respectively. There are many different techniques for generating clock signals C0, C1, C2, and C3. Clock signals C0, C1, C2, and C3 can be generated, for example, by a delay-locked loop, a phase mixer, or a phase-locked loop.

Circuit blocks 511-514 each have 4 multiplexers 571-574 that receive 8 sampled signals Q1-Q8 from SR flip-flops 561-568, respectively, in a corresponding one of circuit blocks 502-505. The output signals of multiplexers 571-574 are M1-M4, respectively.

Circuit block 511 and multiplexers 521-523 select one of the sampled signals from circuit block 502 as sampled bit D1S based on the values of previously sampled bits D2, D3, and D4. Flip-flop 541 stores the value of D1S as sampled bit D1 in response to each rising edge in clock signal CC. Circuit block 512 and multiplexers 524-526 select one of the sampled signals from circuit block 503 as sampled bit D2S based on the values of D3, D4, and D1S. Flip-flop 542 stores the value of D2S as sampled bit D2 in response to each rising edge in clock signal CB.

Circuit block 513 and multiplexers 527-529 select one of the sampled signals from circuit block 504 as sampled bit D3S based on the values of D4, D1S, and D2S. Flip-flop 543 stores the value of D3S as sampled bit D3 in response to each rising edge in clock signal CA. Circuit block 514 and multiplexers 530-532 select one of the sampled signals from circuit block 505 as sampled bit D4S based on the values of D1S, D2S, and D3S. Flip-flop 544 stores the value of D4S as sampled bit D4 in response to each rising edge in clock signal C3.

FIG. 5D is a timing diagram that illustrates example waveforms for clock signals C0, C1, C2, C3, CA, CB, and CC. In the example of FIG. 5D, each of the clock signals has a 50% duty cycle. In alternative embodiments, the clock signals shown in FIG. 5D have duty cycles that are greater than or less than 50%.

As shown in FIG. 5D, clock signals C0, C1, C2, and C3 are equally spaced apart in phase at 90° phase intervals. Clock buffer 551 delays clock signal CC by a delay of γ3 relative to the phase of clock signal C0. Clock buffer 552 delays clock signal CB by a delay of γ2 relative to the phase of clock signal C1. Clock buffer 553 delays clock signal CA by a delay of γ1 relative to the phase of clock signal C2. The relative phases of CA, CB, and CC can be shifted as shown in FIG. 5D without changing the clock periods of these signals.

In one embodiment, γ1=T_(F)/4, γ2=T_(F)/2, γ3=3T_(F)/4, and T_(F)=T_(C,Q)+T_(SU) for each of flip-flops 541-544. In this embodiment, PRE circuit 500 is subject to the timing constraints shown below in equations (4)-(7).

T _(MUX,SO) +T _(F)<1 (UI)+3T _(F)/4   (4)

(2×T _(MUX,SO))+T _(F)<2 (UI)+T _(F)/2   (5)

(3×T _(MUX,SO))+T _(F)<3 (UI)+T _(F)/4   (6)

(4×T _(MUX,SO))+T _(F)<4 (UI)   (7)

Equation (4) can be reduced to T_(MUX)+T_(F)/4<1 (UI), which indicates a substantial reduction in this timing constraint. Equation (4) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(SU) of flip-flop 541 that occurs between the rising edge of C3 that causes flip-flop 544 to update D4 and the next rising edge of CC that causes flip-flop 541 to update D1. PRE circuit 500 implements time borrowing. The delay between each rising edge in C3 and the next rising edge of CC is increased by 3T_(F)/4 to provide additional time to complete the delay T_(C,Q)+T_(MUX,SO)+T_(SU) through flip-flop 544, multiplexer 523, and flip-flop 541, as shown in equation (4).

The delay between each rising edge of CC and the next rising edge of CB is reduced by T_(F)/4, the delay between each rising edge of CB and the next rising edge of CA is reduced by T_(F)/4, and the delay between each rising edge of CA and the next rising edge of C3 is reduced by T_(F)/4. The delays between these rising edges of the clock signals can be reduced without adversely affecting the operation of PRE circuit 500, because the output signals of multiplexers 523, 526, and 529 are transmitted directly to the select inputs of multiplexers 526, 529, and 532, respectively.

Equation (5) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the delay T_(MUX,SO) of multiplexer 526, plus the T_(SU) of flip-flop 542 that occurs between the rising edge of C3 that causes flip-flop 544 to update D4 and the next rising edge of CB that causes flip-flop 542 to update D2. The delay between each rising edge in C3 and the next rising edge of CB is increased by T_(F)/2 to provide additional time to complete the delays through these circuits.

Equation (6) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(MUX,SO) of multiplexer 526, plus the T_(MUX,SO) of multiplexer 529, plus the T_(SU) of flip-flop 543 that occurs between the rising edge of C3 that causes flip-flop 544 to update D4 and the next rising edge of CA that causes flip-flop 543 to update D3. The delay between each rising edge in C3 and the next rising edge of CA is increased by T_(F)/4 to provide additional time to complete the delays through these circuits.

Equation (7) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(MUX,SO) of multiplexer 526, plus the T_(MUX,SO) of multiplexer 529, plus the T_(MUX,SO) of multiplexer 532, plus the T_(SU) of flip-flop 544 that occurs within each period of C3 (i.e., within 4 UIs). The longest signal path delay in PRE circuit 500 is shown in equation (3), where N=4.

FIG. 6A illustrates another example of a partial response equalizer (PRE) circuit 600 that uses a time borrowing technique and that generates each sampled bit based on the values of three previously sampled bits. PRE circuit 600 is also a 3-tap PRE circuit. PRE circuit 600 samples an input signal that has a quad data rate (QDR) with respect to the sampling clock signal.

PRE circuit 600 includes the same circuit elements as PRE circuit 500, except clock buffers 551-553. Instead, PRE circuit 600 has a clock buffer 601. The clock signals used to clock flip-flops 541-544 in PRE circuit 600 have different phase relationships than the clock signals that clock flip-flops 541-544 in PRE circuit 500. In PRE circuit 600, clock signal CX clocks flip-flop 541, clock signal C1 clocks flip-flop 542, clock signal C2 clocks flip-flop 543, and clock signal C3 clocks flip-flop 544.

Clock buffer 601 generates clock signal CX in response to clock signal C0. FIG. 6B is a timing diagram that illustrates example waveforms for clock signals C0, C1, C2, C3, and CX. As shown in FIG. 6B, clock buffer 601 provides a delay of γ to clock signal CX relative to the phase of clock signal C0.

In one embodiment, γ=T_(F)/2, and T_(F)=T_(C,Q)+T_(SU) for each of flip-flops 541-544. In this embodiment, PRE circuit 600 is subject to the timing constraints shown below in equations (8)-(11).

T _(MUX,SO) +T _(F)<1 (UI)+T _(F)/2   (8)

(2×T _(MUX,SO))+T _(F)<2 (UI)   (9)

(3×T _(MUX,SO))+T _(F)<3 (UI)   (10)

(4×T _(MUX,SO))+T _(F)<4 (UI)   (11)

Equation (8) can be reduced to T_(MUX)+T_(F)/2 <1 (UI). Equation (8) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(SU) of flip-flop 541 that occurs between the rising edge of C3 that causes flip-flop 544 to update D4 and the next rising edge of CX that causes flip-flop 541 to update D1. PRE circuit 600 also implements time borrowing. The delay between each rising edge in C3 and the next rising edge of CX is increased by T_(F)/2 to provide additional time to complete the delay T_(C,Q)+T_(MUX,SO)+T_(SU) through flip-flop 544, multiplexer 523, and flip-flop 541, as shown in equation (8). The delay between each rising edge in CX and the next rising edge in C1 is reduced by T_(F)/2.

Equation (9) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(MUX,SO) of multiplexer 526, plus the T_(SU) of flip-flop 542 that occurs between each rising edge of C3 and the next rising edge of C1. Equation (10) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(MUX,SO) of multiplexer 526, plus the T_(MUX,SO) of multiplexer 529, plus the T_(SU) of flip-flop 543 that occurs between each rising edge of C3 and the next rising edge of C2. Equation (11) represents the timing constraint for the T_(C,Q) of flip-flop 544, plus the T_(MUX,SO) of multiplexer 523, plus the T_(MUX,SO) of multiplexer 526, plus the T_(MUX,SO) of multiplexer 529, plus the T_(MUX,SO) of multiplexer 532, plus the T_(SU) of flip-flop 544 that occurs within each period of C3.

In alternative embodiments of PRE circuits 500 and 600, non-sequential logic circuits (e.g., inverters) are coupled between the outputs of multiplexers 523, 526, and 529 and the select inputs of multiplexers 526, 529, and 532 in the transmit paths of signals D1S, D2S, and D3S, respectively.

In other embodiments of the PRE circuits described herein, the flip-flops can be substituted with latches or another suitable type of storage circuit. Suitable storage circuits may either rely on static or dynamic storage techniques, which may store logic values temporarily by charging or discharging a capacitor.

A receiver containing one of the partial response equalizers described herein can be used in any suitable integrated circuit, such as, for example, a memory integrated circuit or a controller integrated circuit. The receiver containing the partial response equalizer can, e.g., be used to sample input data received from an external source before the data is stored in memory cells inside the memory integrated circuit (e.g., a DRAM or SRAM chip).

The teachings provided herein can be readily extended to additional systems that rely upon sampler circuits using multiple thresholds, such as 4-PAM systems. PAM refers to pulse amplitude. The design for additional embodiments having multiple thresholds is generally similar to the embodiment described herein. These additional embodiments can involve additional sampler paths and multi-bit connections between the multiplexers.

The embodiments shown in the figures and described herein can be implemented in hardwired electronic circuitry, in software code stored on a computer readable medium, or using a combination of hardwired circuitry and software.

The foregoing description of the exemplary embodiments has been presented for the purposes of illustration and description. The foregoing description is not intended to be exhaustive or limiting to the examples disclosed herein. In some instances, features described herein can be employed without a corresponding use of other features as set forth. Many modifications, substitutions, and variations are possible in light of the above teachings. 

1. (canceled)
 2. An integrated circuit comprising: a two phase partial response equalizer circuit; the two phase partial response equalizer circuit having a first sampler circuit to sample an input signal to generate a first sampled signal in response to a first sampling clock having a first phase, and a second sampler circuit to sample the input signal to generate a second sampled signal in response to a second sampling clock having a second phase; the two phase partial response equalizer circuit having a first feedback path to control partial response selection by the first sampler circuit in dependence on the second sampled signal and a second feedback path to control partial response selection by the second sampler circuit in dependence on the first sampled signal; a latch in the first feedback path, the second feedback path not having a latch; and circuitry to generate a latch clock for the latch in the first feedback path to be phase offset from each of the first sampling clock and the second sampling clock.
 3. The integrated circuit of claim 2, wherein: the circuitry to generate the latch clock for the latch comprises a series of delay elements to generate the latch clock for the latch from one of the first sampling clock or the second sampling clock.
 4. The integrated circuit of claim 2, wherein: the circuitry to generate the latch clock for the latch comprises a locked loop circuit to generate the latch clock for the latch.
 5. The integrated circuit of claim 2, wherein: the circuitry to generate the latch clock for the latch comprises a phase mixer to generate the latch clock for the latch.
 6. The integrated circuit of claim 2, wherein: the two phase partial response equalizer circuit comprises plural taps to equalize the input signal for each phase in dependence on respective prior bits conveyed by the input signal.
 7. The integrated circuit of claim 2, wherein: the latch in the first feedback path is a first latch, the first latch to latch output of the first sampled signal; the latch clock for the latch is a first latch clock for the first latch; the integrated circuit further comprises a second latch to latch output of the second sampled signal; and the circuitry is further operable to generate a second latch clock for the second latch by delaying the first latch clock by an amount approximately equal to a unit interval plus one-half of a delay time of the first latch.
 8. The integrated circuit of claim 7, wherein: the delay time of the first latch is approximately equal to the sum of a clock-to-Q time plus a setup time of the first latch.
 9. An integrated circuit comprising: a multiphase partial response equalizer circuit; the multiphase partial response equalizer circuit having sampler circuits to sample an input signal to generate respective sampled signals in response to sampling clocks having respective phases; the multiphase partial response equalizer circuit having a feedback path for each of the sampler circuits to control partial response selection in dependence on the sampled signals from another of the sampler circuits; a latch in the feedback path for a first one of the sampler circuits, the feedback path for at least a second one of the sampler circuits not having a latch; circuitry to generate a latch clock for the latch in the feedback path so as to have a phase offset from each of the sampling clocks.
 10. The integrated circuit of claim 9, wherein: the multiphase partial response equalizer circuit is a two phase partial response equalizer circuit; the sampler circuits include a first sampler circuit and a second sampler circuit, the sampling clocks include a first sampling clock and a second sampling clock, the feedback paths include a first feedback path and a second feedback path, and the sampled signals include a first sampled signal and a second sampled signal; the first sampler circuit is to sample the input signal to generate the first sampled signal in response to the first sampling clock, and the second sampler circuit is to sample the input signal to generate the second sampled signal in response to the second sampling clock; the first feedback path is to control partial response selection by the first sampler circuit in dependence on the second sampled signal and the second feedback path is to control partial response selection by the second sampler circuit in dependence on the first sampled signal; the latch is in the first feedback path; and the circuitry is to generate the latch clock for the latch in the first feedback path so as to be phase offset from each of the first sampling clock and the second sampling clock.
 11. The integrated circuit of claim 9, wherein: the circuitry comprises a series of delay elements to generate the latch clock for the latch from one of the sampling clocks.
 12. The integrated circuit of claim 9, wherein: the latch in the feedback path of the first one of the sampler circuits is a first latch, the first latch to latch one of the sampled signals output by the first one of the sampler circuits; the latch clock for the latch is a first latch clock for the first latch; the integrated circuit further comprises a second latch to latch output of another of the sampled signals output by the second one of the sampler circuits; and the circuitry is further operable to generate a second latch clock for the second latch by delaying the first latch clock.
 13. The integrated circuit of claim 12, wherein the circuitry is to generate the second latch clock from the first latch clock by delaying the first latch clock by an amount approximately equal to a unit interval plus one-half of a delay time of the first latch.
 14. The integrated circuit of claim 9, wherein the multiphase partial response equalizer circuit is a four phase partial response equalizer circuit.
 15. The integrated circuit of claim 14, wherein: the sampler circuits include a first sampler circuit, a second sampler circuit, a third sampler circuit and a fourth sampler circuit, the sampling clocks include a first sampling clock, a second sampling clock, a third sampling clock and a fourth sampling clock, the feedback paths include a first feedback path, a second feedback path, a third feedback path and a fourth feedback path, and the sampled signals include a first sampled signal, a second sampled signal, a third sampled signal and a fourth sampled signal; the first sampler circuit is to sample the input signal to generate the first sampled signal in response to the first sampling clock, the second sampler circuit is to sample the input signal to generate the second sampled signal in response to the second sampling clock, the third sampler circuit is to sample the input signal to generate the third sampled signal in response to the third sampling clock, the fourth sampler circuit is to sample the input signal to generate the fourth sampled signal in response to the fourth sampling clock; the first feedback path is to control partial response selection by the first sampler circuit in dependence on the fourth sampled signal, the second feedback path is to control partial response selection by the second sampler circuit in dependence on the first sampled signal, the third feedback path is to control partial response selection by the third sampler circuit in dependence on the second sampled signal, and the fourth feedback path is to control partial response selection by the fourth sampler circuit in dependence on the third sampled signal; the latch is in the first feedback path, and no latch is in any of the second feedback path, the third feedback path and the fourth feedback path; and the circuitry is to generate the latch clock for the latch in the first feedback path so as to be phase offset from each of the first sampling clock, the second sampling clock, the third sampling clock and the fourth sampling clock.
 16. The integrated circuit of claim 15, wherein: the latch in the first feedback path is a first latch, the first latch to latch output of the first sampled signal; the latch clock for the latch is a first latch clock for the first latch; the integrated circuit further comprises a second latch to latch output of the second sampled signal, a third latch to latch output of the third sampled signal and a fourth latch to latch output of the fourth sampled signal; and the circuitry is further operable to generate a second latch clock for the second latch by delaying the first latch clock, a third latch clock for the third latch by delaying the second latch clock, and a fourth latch clock for the fourth latch by delaying the third latch clock.
 17. The integrated circuit of claim 16, where the delay of the second latch clock relative to the first latch clock, the delay of the third latch clock relative to the second latch clock, and the delay of the third latch clock relative to the fourth latch clock is each approximately equal to a unit interval plus one-quarter of a delay time of the first latch.
 18. An integrated circuit comprising: a multiphase partial response equalizer circuit; the multiphase partial response equalizer circuit having sampler circuits to sample an input signal to generate respective sampled signals in response to sampling clocks having respective phases, and a latch for each sampled signal to latch output of the respective sampled signal; the multiphase partial response equalizer circuit having a feedback path for each of the sampler circuits to control partial response selection in dependence on one of the sampled signals from another of the sampler circuits; exactly one of the feedback paths having a latch of the latches for the respective sampled signals; means for utilizing time borrowing to generate a latch clock for each respective latch, so as to spread a delay time associated with the latch in the exactly one of the feedback paths across the latch clocks, at least one of the latch clocks being phase-offset relative to all of the sampling clocks.
 19. A method comprising: providing a multiphase partial response equalizer circuit to sample an input signal, including a sampler circuit for each of multiple phases of the input signal to generate a respective sampled signal and a latch for each sampled signal to latch output of the respective sampled signal; generating sampling clocks for respective ones of the sampler circuits, the sampling clocks having respective phases; providing a feedback path for each of the sampler circuits to control partial response selection in dependence on one of the sampled signals from another of the sampler circuits, wherein exactly one of the feedback paths has a latch of the latches for the respective sampled signals; and providing circuitry to generate a latch clock for each respective latch using time borrowing, so as to spread a delay time associated with the latch in the exactly one of the feedback paths across the latch clocks, at least one of the latch clocks being phase-offset relative to all of the sampling clocks.
 20. The method of claim 19, where providing circuitry comprises fixing a time offset for each of the latch clocks, so to deliberately offset each latch clock relative to another of the latch clocks to spread the delay time.
 21. The method of claim 20, where the time offset for each latch clock is approximately equal to a sum of the delay time associated with the latch divided by the number of phases of the multiple phases plus a unit interval. 